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SC4524B
16V 2A Step-Down Switching Regulator
POWER MANAGEMENT Features

Description
The SC4524B is a constant frequency peak current-mode step-down switching regulator capable of producing 2A output current from an input ranging from 3V to 16V. The switching frequency of the SC4524B is programmable up to 2MHz, allowing the use of small inductors and ceramic capacitors for miniaturization, and high input/ output conversion ratio. The SC4524B is suitable for next generation XDSL modems, high-definition TVs and various point of load applications. Peak current-mode PWM control employed in the SC4524B achieves fast transient response with simple loop compensation. Cycle-by-cycle current limiting and hiccup overload protection reduces power dissipation during output overload. Soft-start function reduces input startup current and prevents the output from overshooting during power-up. The SC4524B is available in SOIC-8 EDP package.
Wide input range: 3V to 16V 2A Output Current 200kHz to 2MHz Programmable Frequency Precision 1V Feedback Voltage Peak Current-Mode Control Cycle-by-Cycle Current Limiting Hiccup Overload Protection with Frequency Foldback Soft-Start and Enable Thermal Shutdown Thermally Enhanced 8-pin SOIC Package Fully RoHS and WEEE compliant
Applications

XDSL and Cable Modems Set Top Boxes Point of Load Applications CPE Equipment DSP Power Supplies LCD and Plasma TVs
Typical Application Circuit
10V - 16V C4 2.2mF D1 1N4148 C1 0.1mF L1 6.8mH R4 102k
SS270 REV 4
Efficiency
V IN
90 85
OUT 5V/2A
IN
BST SW
80
Efficiency (%)
75 70 65 60 55 50 45 40
VIN = 12V
SS/EN
SC4524B
FB
COMP C7 10nF R7 30.1k C5 1nF
ROSC
GND D2 20BQ030 R6 25.5k C2 22mF
C8 10pF
R5 18.2k
L1: Wurth 744 778 9006
C2: Murata GRM31CR60J226K C4: Murata GRM31CR61E225K
0
0.5
1
1.5
2
Load Current (A)
Figure 1. 1MHz 10V-16V to 5V/2A Step-down Converter
November 1, 2007 1
SC4524B
Pin Configuration Ordering Information
Device
SC4524BSETRT(1)(2)
SW IN ROSC GND 1 2 3 4 9 8 7 6 5 BST FB COMP SS/EN
Package
SOIC-8 EDP Evaluation Board
SC4524BEVB
Notes: (1) Available in tape and reel only. A reel contains 2,500 devices. (2) Available in lead-free package only. Device is fully WEEE and RoHS compliant.
(8 - Pin SOIC - EDP)
Marking Information
yyww=Date code (Example: 0752) xxxxx=Semtech Lot No. (Example: E9010)
2
SC4524B
Absolute Maximum Ratings
VIN Supply Voltage .................................... -0.3 to 24V BST Voltage ...................................................... 40V BST Voltage above SW .......................................... 24V SS Voltage ...................................................-0.3 to 3V FB Voltage ................................................... -0.3 to VIN SW Voltage ................................................ -0.6 to VIN SW Transient Spikes (10ns Duration)......... -2.5V to VIN +1.5V Peak IR Reflow Temperature ............................... 260C ESD Protection Level(2) ....................................... 2000V
Thermal Information
Junction to Ambient (1) .................................... 36C/W Junction to Case (1) ....................................... 5.5C/W Maximum Junction Temperature........................... 150C Storage Temperature .............................. -65 to +150C Lead Temperature (Soldering) 10 sec ..................... 300C
Recommended Operating Conditions
Input Voltage Range .................................... 3V to 16V Maximum Output Current .................................... 2A
Exceeding the above specifications may result in permanent damage to the device or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not recommended. NOTES(1) Calculated from package in still air, mounted to 3" x 4.5", 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards. (2) Tested according to JEDEC standard JESD22-A114-B.
Electrical Characteristics
Unless otherwise noted, VIN = 12V, VBST = 15V, VSS = 2.2V, -40C < TA = TJ < 125C, ROSC = 12.1k.
Parameter Input Supply
Input Voltage Range VIN Start Voltage VIN Start Hysteresis VIN Quiescent Current VIN Quiescent Current in Shutdown
Conditions
Min
3
Typ
Max
16
Units
V V mV
VIN Rising
2.70
2.82 225
2.95
VCOMP = 0 (Not Switching) VSS/EN = 0, VIN = 12V
2 40
2.6 50
mA A
Error Amplifier
Feedback Voltage Feedback Voltage Line Regulation FB Pin Input Bias Current Error Amplifier Transconductance Error Amplifier Open-loop Gain COMP Pin to Switch Current Gain COMP Maximum Voltage COMP Source Current COMP Sink Current VIN = 3V to 16V VFB = 1V, VCOMP = 0.8V 0.980 1.000 0.005 -170 280 60 8 -340 1.020
V %/V nA
-1
dB A/V
V A
VFB = 0.9V VFB = 0.8V, VCOMP = 0.8V VFB = 1.2V, VCOMP = 0.8V
(Note 1) ISW = -2.6A 2.6
2.4 17 25
Internal Power Switch
Switch Current Limit Switch Saturation Voltage 3.3 250 4.3 400 A mV
3
SC4524B
Electrical Characteristics (Cont.)
Unless otherwise noted, VIN = 12V, VBST = 15V, VSS = 2.2V, -40C < TA = TJ < 125C, ROSC = 12.1k.
Parameter
Minimum Switch On-time Minimum Switch Off-time Switch Leakage Current Minimum Bootstrap Voltage BST Pin Current
Conditions
Min
Typ
135 100
Max
150 10
Units
ns ns A V mA
ISW = -2.6A ISW = -2.6A
1.8 60
2.3 95
Oscillator
Switching Frequency ROSC = 12.1k ROSC = 93.1k ROSC = 12.1k, VFB = 0 ROSC = 93.1k, VFB = 0 1.04 240 110 50 1.3 300 230 100 1.56 360 350 170 MHz kHz kHz
Foldback Frequency
Soft Start and Overload Protection
SS/EN Shutdown Threshold SS/EN Switching Threshold Soft-start Charging Current Soft-start Discharging Current Hiccup Arming SS/EN Voltage Hiccup SS/EN Overload Threshold Hiccup Retry SS/EN Voltage 0.2 0.3 1.13 1.7 1.2 2.0 1.5 2.8 0.4 1.3 V V A A V V 1.2 V
VFB = 0 V VSS/EN = 0 V VSS/EN = 1.5 V VSS/EN Rising VSS/EN Falling VSS/EN Falling
1.0
2.15 1.9 0.6 1.0
Over Temperature Protection
Thermal Shutdown Temperature Thermal Shutdown Hysteresis Note 1: Switch current limit does not vary with duty cycle. 165 10 C C
4
SC4524B
Pin Descriptions
SO-8
1 2 3 4
Pin Name
SW IN ROSC GND
Pin Function
Emitter of the internal NPN power transistor. Connect this pin to the inductor, the freewheeling diode and the bootstrap capacitor. Power supply to the regulator. It is also the collector of the internal NPN power transistor. It must be closely bypassed to the ground plane. An external resistor from this pin to ground sets the oscillator frequency. Ground pin Soft-start and regulator enable pin. A capacitor from this pin to ground provides soft-start and overload hiccup functions. Hiccup can be disabled by overcoming the internal soft-start discharging current with an external pullup resistor connected between the SS/EN and the IN pins. Pulling the SS/EN pin below 0.2V completely shuts off the regulator to low current state. The output of the internal error amplifier. The voltage at this pin controls the peak switch current. A RC compensation network at this pin stabilizes the regulator. The inverting input of the error amplifier. If VFB falls below 0.8V, then the switching frequency will be reduced to improve short-circuit robustness (see Applications Information for details). Supply pin to the power transistor driver. Tie to an external diode-capacitor bootstrap circuit to generate drive voltage higher than VIN in order to fully enhance the internal NPN power transistor. The exposed pad serves as a thermal contact to the circuit board. It is to be soldered to the ground plane of the PC board.
5
SS/EN
6 7 8 9
COMP FB BST Exposed Pad
5
SC4524B
Block Diagram
IN COMP
6
SLOPE COMP
2
+
+ EA +
S
+
+ ISEN 6.1mW OC + ILIM 20mV BST
FB
7
V1 + PWM FREQUENCY FOLDBACK
8
S R CLK Q POWER TRANSISTOR
ROSC
3
OSCILLATOR
R R SS/EN
5
OVERLOAD
A1
+ -
1.23V
1
SW
PWM
1
GND
4
1V
1.9V FAULT
REFERENCE & THERMAL SHUTDOWN
SOFT-START AND OVERLOAD HICCUP CONTROL
Figure 2. SC4524B Block Diagram
1.9V IC 2mA
B4 + B1
S Q R OVERLOAD
SS/EN
1V/2.15V
B2
FAULT
ID 3.5mA
_ Q
S R
OC
PWM
B3
Figure 3. Soft-start and Overload Hiccup Control Circuit
6
(2)
SC4524A/B
(3)
SC4524A/B SS270 REV 6-7
SC4524B
Typical Characteristics
Efficiency
V O=5V V O=3.3V V O=2.5V
Efficiency
90 85 80 75 70 65 60 55 50 40
V O=3.3V V O=2.5V V O=1.5V
Feedback Voltage vs Temperature
1.02 V IN =12V 1.01 1.00 0.99 0.98 0.97
90 85 80 75 70 65 60 55 50 40 0 SS270 REV 6-7 0.5
Efficiency (%)
Efficiency (%)
V O=1.5V
VO=1.0V
(5) 45
1MHz, VIN =12V D 2 =20BQ030
(6) 45
1.5 2 0 0.5 SS270 REV 6-7
1MHz, VIN=5V D2 =20BQ030
1
1
1.5
2
VFB (V)
-50
-25
0
25
50
75
o
100 125
Load Current (A)
Load Current (A)
Temperature ( C)
1000
Frequency Setting Resistor vs Frequency
V IN =12V
Frequency vs Temperature
1.2
R OSC=93.1k
Foldback Frequency vs VFB
1.25 1
ROSC=93.1k
1.1
100
Normalized Frequency
Normalized Frequency
ROSC (k)
0.75 0.5 0.25
R OSC=12.1k
1.0
R OSC=12.1k
10
(8)
1 0 0.5 1 1.5 2 2.5
0.9
(9)
-50 -25 0 25 50 75 100 125 Temperature (o C)
TA =25oC
0.8
0 0.0 0.2 0.4 0.6 0.8 1.0
Frequency (MHz)
SS270 REV 6-7 SS270 REV 6-7
VFB (V)
SS270 REV 6-7
300 250 200 150 100 50 0.0
Switch Saturation Voltage vs Switch Current
Switch Current Limit vs Temperature
4.5 100.0
BST Pin Current vs Switch Current
V IN =12V
-40oC 125oC 25oC
BST Pin Current (mA)
Current Limit (A)
4.0
75.0
V BST =15V
V CESAT (mV)
3.5
50.0
-40oC 125oC
3.0
25.0
2.5 0.5 1.0 1.5 2.0 2.5 -50 -25 0 25 50
o
0.0 75 100 125 0 0.5 1 1.5 2 2.5 3 Temperature ( C) Switch Current (A)
Switch Current (A)
7
(11)
SS270 REV 6-7
(12)
SS270 REV 6-7
SC4524B
Typical Characteristics (Cont.)
SS270 REV 6-7
VIN Thresholds vs Temperature
3.0 2.9
Start
2.5 2.0
VIN Supply Current vs Soft-Start Voltage
125oC -40oC
50 40
Current (uA)
VIN Shutdown Current vs VIN
V SS = 0
VIN Threshold (V)
Current (mA)
2.8 2.7 2.6
1.5 1.0 0.5 0.0
30 20 10 0
-40oC
125oC
(14) 2.5
2.4 -50 -25 0
UVLO
(15)
0 0.5 1 V SS (V) 1.5 2
SS270 REV 6-7
25
50
75
100 125
0
2
4
6
8
VIN (V)
10
12
14
16
Temperature (o C) SS270 REV 6-7 SS270 REV 6-7
VIN Quiescent Current vs VIN
2.5 125oC 2.0
0.40
SS Shutdown Threshold vs Temperature
Soft-Start Charging Current vs Soft-Start Voltage
0.0 -0.5
SS Threshold (V)
Current (mA)
Current (uA)
-40oC 1.5 1.0 0.5 VCOMP = 0 0.0 0 2 4 6 8 V IN (V) 10 12 14 16
0.35 -1.0 -1.5 -2.0 -2.5 0.20 -50 -25 0 25 50
o
125oC
0.30
-40oC
0.25
-3.0 75 100 125 0 0.5 1 V SS (V) 1.5 2
Temperature ( C)
8
SC4524B
Applications Information
Operation The SC4524B is a constant-frequency, peak current-mode, step-down switching regulator with an integrated 16V, 2.6A power NPN transistor. Programmable switching frequency makes the regulator design more flexible. With the peak current-mode control, the double reactive poles of the output LC filter are reduced to a single real pole by the inner current loop. This simplifies loop compensation and achieves fast transient response with a simple Type-2 compensation network. As shown in Figure 2, the switch collector current is sensed with an integrated 6.1mW sense resistor. The sensed current is summed with a slope-compensating ramp before it is compared with the transconductance error amplifier (EA) output. The PWM comparator trip point determines the switch turn-on pulse width. The current-limit comparator ILIM turns off the power switch when the sensed signal exceeds the 20mV current-limit threshold. Driving the base of the power transistor above the input power supply rail minimizes the power transistor saturation voltage and maximizes efficiency. An external bootstrap circuit (formed by the capacitor C1 and the diode D1 in Figure 1) generates such a voltage at the BST pin for driving the power transistor. Shutdown and Soft-Start The SS/EN pin is a multiple-function pin. An external capacitor (4.7nF to 22nF) connected from the SS pin to ground sets the soft-start and overload shutoff times of the regulator (Figure 3). The effect of VSS/EN on the SC4524B is summarized in Table 1. Table 1: SS/EN operation modes
SS/EN <0.2V SS/EN to 1.23V 0.4V <0.2V 1.23V to 2.1V Mode Supply Current Shutdown 18uA @ 5Vin Mode switching Supply Current Not 2mA Shutdown 18uA @ 5Vin Switching & hiccup disabled Load dependent Not switching hiccup armed 2mA Switching & Load dependent
When the SS/EN pin is released, the soft-start capacitor is charged with an internal 1.6A current source (not shown in Figure 3). As the SS/EN voltage exceeds 0.4V, the internal bias circuit of the SC4524B turns on and the SC4524B draws 2mA from VIN. The 1.6A charging current turns off and the 2A current source IC in Figure 3 slowly charges the soft-start capacitor. The error amplifier EA in Figure 2 has two non-inverting inputs. The non-inverting input with the lower voltage predominates. One of the non-inverting inputs is biased to a precision 1V reference and the other non-inverting input is tied to the output of the amplifier A1. Amplifier A1 produces an output V1 = 2(VSS/EN -1.23V). V1 is zero and COMP is forced low when VSS/EN is below 1.23V. During start up, the effective non-inverting input of EA stays at zero until the soft-start capacitor is charged above 1.23V. Once VSS/EN exceeds 1.23V, COMP is released. The regulator starts to switch when VCOMP rises above 0.4V. If the soft-start interval is made sufficiently long, then the FB voltage (hence the output voltage) will track V1 during start up. VSS/EN must be at least 1.83V for the output to achieve regulation. Proper soft-start prevents output overshoot. Current drawn from the input supply is also well controlled. Overload / Short-Circuit Protection Table 2 lists various fault conditions and their corresponding protection schemes in the SC4524B. Table 2: Fault conditions and protections Cycle-by-cycle limit at
Over current Fault Protective Action frequency programmed Cycle-by-cycle limit at limit with Cycle-by-cycle Condition Fault Protective Action
IL>ILimit, VFB>0.8V Condition
IL>ILimit, VFB>0.8V Over current current IL>ILimit, VFB<0.8V Over
programmed frequency frequency foldback Cycle-by-cycle limit with retry VSS/EN Falling Persistent over current Shutdown, then IL>ILimit, VFB<0.8V Over current SS/EN<1.9V or short circuit frequency foldback (Hiccup) VSS/EN Falling Persistent over current Shutdown, then retry Tj>160C Over temperature Shutdown SS/EN<1.9V or short circuit (Hiccup) Tj>160C Over temperature Shutdown
0.4V to 1.23V >2.1V 1.23V to 2.1V >2.1V
Switching & hiccup disabled Switching & hiccup armed
Pulling the SS/EN pin below 0.2V shuts off the regulator and reduces the input supply current to 18A (VIN = 5V).
As summarized in Table 1, overload shutdown is disabled during soft-start (VSS/EN<2.1V). In Figure 3, the reset input of the overload latch B2 will remain high if the SS/EN voltage is below 2.1V. Once the soft-start capacitor is charged above 2.1V, the output of the Schmitt trigger B1 goes high, the reset input of B2 goes low and hiccup becomes armed.
9
SC4524B
Applications Information (Cont.)
As the load draws more current from the regulator, the current-limit comparator ILIM (Figure 2) will eventually limit the switch current on a cycle-by-cycle basis. The over-current signal OC goes high, setting the latch B3. The soft-start capacitor is discharged with (ID - IC) (Figure 3). If the inductor current falls below the current limit and the PWM comparator instead turns off the switch, then latch B3 will be reset and IC will recharge the soft-start capacitor. If over-current condition persists or OC becomes asserted more often than PWM over a period of time, then the soft-start capacitor will be discharged below 1.9V. At this juncture, comparator B4 sets the overload latch B2. The soft-start capacitor will be continuously discharged with (ID - IC). The COMP pin is immediately pulled to ground. The switching regulator is shut off until the soft-start capacitor is discharged below 1.0V. At this moment, the overload latch is reset. The soft-start capacitor is recharged and the converter again undergoes soft-start. The regulator will go through soft-start, overload shutdown and restart until it is no longer overloaded.
AC = V R4 = R6 O - 1 down switching Vregulator in continuous-conduction 1.0 mode (CCM) is given by D= VO + VD VIN + VD - VCESAT
(2)
AC =
where VCESAT is the switch saturation voltage and VD is voltage drop across the rectifying diode. ( V + VD ) (1 - D) DIL = O FSW L In peak current-mode1 control, the PWM modulating ramp is the sensed current ramp of the power switch. This currentVrampD is (absent unless the switch is turned ( + V ) 1 - D) L= O on. The1intersection ofSW ramp with the output of the 20% IO F this voltage feedback error amplifier determines the switch pulse width. The propagation delay time required to immediately turn off D (1 - D) after it is turned on is the IRMS _ CIN = IO the switch minimum controllable switch on time (TON(MIN)). Closed-loop measurement shows that the SC4524B 1 minimum onDtimeESR + DVO = IL is about 135ns at room temperature FSW on (Figure 4). If the required8switch C O time is shorter than the minimum on time, the regulator will either skip cycles or it will jitter.
SS270 REV 6-7
R7 = C5 = C8 =
Vo = Vc
Fig.4
GPWM
If the FB voltage falls below 0.8V because of output overload, then the switching frequency will be reduced. Frequency foldback helps to limit the inductor current when the output is hard shorted to ground. During normal operation, the soft-start capacitor is charged to 2.4V.
R7 = C5 = C8 =
C IN >
IO Minimum On 4 DVIN FSW Time vs Temperature
200 190 180 170 V O =1.5V 1MHz
TON(MIN) (ns)
Setting the Output Voltage The regulator output voltage is set with an external resistive divider (Figure 1) with its center tap tied to the FB pin. For a given R6 value, R4 can be found by
160 150 140 130
V R4 = R6 O - 1 1.0 V
120 1 VFB 1 A C = - 20 log 110 G R 2F C V CO O CA S 100 -50 -25 0 25 50 75 100 125
(1)
VO + VD Setting the Switching Frequency D= VIN + VD - VCESAT The switching frequency of the SC4524B is set with an external resistor from the ROSC pin to ground. ( V + VD ) (1 - D) DIL On O Minimum = Time Consideration FSW L 1
The operating duty cycle of a non-synchronous step( V + VD ) (1 - D) L1 = O 20% IO FSW
1 Temperature (OC) 1 1.0 A C = - 20 log = 15 -3 3 -6 3 .3 2 80 10 22 10 28 6.1 10 Figure 4. Variation of Minimum On Time with Ambient Temperature 15.9 10 20 R7 = = 22.3k 0.28 10 -3 To allow for transient headroom, the minimum operating switch on time should be at least 20% to 30% higher than 1 C5 = = 0.45nF 3 the worst-case minimum10 3time. 2 16 10 22.1 on C8 = 1 = 12pF 2 600 10 3 22.1 10 3
10
L1 =
20% IO FSW
O
D
SC4524B
IRMS _ CIN = IO D (1 - D)
Vo = Vc
Applications Information (Cont.)
Minimum Off Time Limitation The PWM latch in Figure 2 is reset every cycle by the clock. The clock also turns off the power transistor to refresh the bootstrap capacitor. This minimum off time limits the attainable duty cycle of the regulator at a given switching frequency. The measured minimum off time is 100ns typically. If the required duty cycle is higher than the attainable maximum, then the output voltage will not be able toR6 VOits-set value in continuous-conduction R4 = reach 1 1.0 V mode. Inductor Selection D VO + V D= VIN + VD - VCESAT The inductor ripple current for a non-synchronous stepdown converter in continuous-conduction mode is
1 DV = DIL ESR + The inputOcapacitance must also be high enough to keep 8 FSW C O within specification. This is important input ripple voltage in reducing the conductive EMI from the regulator. The input capacitance can be estimated from
GPWM
R7 = AC =
( V V V ) (1 - D) + D R R4IL== 6 O O D - 1 FV 1.0SW L 1
(3)
where FSW is the switching frequency and L1 is the ( + +V inductance. VO VOVD ) D(1 - D) L 1== D VIN 20% -OVCESAT + VD I FSW An inductor ripple current between 20% to 50% of the maximum load current gives a good compromise among IRMS CINV= + V size.-- D) ( D) O efficiency,_ cost Iand )D (11 Re-arranging Equation (3) and ( D DIL = O inductor ripple current, the inductor is assuming 35% F L SW 1 given by
( V + V ) (1 - D) 1 L 1 O = DIL ESR + DV = O D (4) 35V IO FSW FSW C O % 8 O -1 R4 = R6 If the input voltage varies over a wide range, then choose 1.0 V L1 based on the nominal input voltage. Always verify IRMS = I D (1 - D) converter _ CIN V O+ Vat the input voltage extremes. operation OI D D= C IN > + V O- V VIN DV F CESAT 4D The peak current IN SW SC4524B power transistor is at limit of least 2.6A. The maximum deliverable load current for the 1 DV = DIL SC4524B Ois 2.6A ESR + one half of the inductor ripple minus 8 F C ( V +VD ) (1 - D)SW O current.IL = O D FSW L 1
Input Decoupling Capacitor ( V + VD ) (1 - D) L 1 = O IO C IN > 20% I should be chosen to handle the RMS The input capacitorO FFSW 4 DVIN SW ripple current of a buck converter. This value is given by
IRMS _ CIN = IO D (1 - D)
(5)
I C R VO (6) C5 = R4IN=> 6 O - 1 41.VIN FSW D0 V AC = V 1 DV log the 1 Awhere20 IN is allowable input ripple voltage. FB C=- G R 2F C C8 = VO +S CO VOCA VD D= Multi-layerVceramic capacitors, which have very low ESR IN + VD - VCESAT 1 1 1 (a few mW) and can easily handle high RMS ripple current,.0 A C = - 20 log == R 15 -3 3 -6 are the ideal choice1for input 80 10 A single 4.7F.3 7 3 28 6. 10 2 filtering. 22 10 X5R ceramic( V + V ) (1 - D) capacitor is adequate for 500kHz or higher O D DIL frequency applications, and 10F is adequate C 5 = switching = 15.9 FSW L 1 for 200kHz to 500kHz switching frequency. For high 10 20 V R7 = =1 .3k 1 22 Avoltage applications, a small ceramic - 20 log-3 FB (1F or 2.2F) can be C = C = 0.28 10 8 R ( VO G CAwith 2F) CESR VO - DC + VD )S (1a low O electrolytic capacitor to placedLin = parallel 1 1 20% I FSW Csatisfy both the3ESROand bulk = 0.45nF requirements. 5= capacitance 2 16 10 22.1 10 3 1 1 1.0 A C = - 20 log = 15 28 6.1 10 - 3 2 80 10 3 22 10 -6 3.3 Vo 1 Output Capacitor = C8 = I = 12pF Vc 2 RMS _ CIN 10O 22. (110D) 600 = I 3 D 1 - 3 The output .9 15 ripple voltage DVO of a buck converter can be 10 20 Rexpressed as -3 = 22.3k 7= GPWM Vo 0.28 10 (1 + sRESR C O ) GPWM = 1 2 Vc (1 DVO/= D)I(1 +ESR + Q + s 2 / n ) + s p L1 s / n (7) 8 3FSW0.C O nF C5 = = 45 3 2 16 10 22.1 10 where CO is the output capacitance. R7 = R 1 1 1 1 1 V,FB GPWM - 20 log, 3 Z = , C8 = = AC = G R p 3 C 12pF R as 2 the S G R ripple current 600 10 1 10 Since CA inductor22.2FC CR O VO DIL increasesESRC OD O CAO S I C IN decreases >(Equation (3)), the output ripple voltage is C 5 = AC 4 DV FSW therefore the IN 1 highest when V is at its maximum. 10 20 1 1.0 R7 = - 20 log (1 + sRESR C O )IN GPWM Vo AC = 15 -3 3 -6 = gm 3 .3 28 6.1 10 2 280 10 22 10 Vc 10F+to / p)(1 + s ceramic s 2 / n ) is found adequate C 8 = (1 s 47F X5R / n Q + capacitor A 1 Cfor=output filtering in most applications. Ripple current 5 2 FZ115.9 R in the10R20 7 capacitor is not a concern because the output 1 1 Rinductor = GPWM 1current= of a3kp converter directly=feeds C ,, , 22. buck , Z 7 0.28 10S-3 GCA R CO R ESRC OO Cresulting in very low rippleRcurrent. Avoid using Z5U 8= 2 FP1 R7 1 Cand Y5VC ceramic capacitors for output filtering because = 0.45nF A 5= 3 2 10 3 22.1 have 10 2016 of capacitors10 high temperature and high Rthese types 7= gm voltage coefficients. 1 C8 = = 12pF 2 600 10 3 22.1 10 3 1 CFreewheeling Diode 5= 2 FZ1 R7 G 1 + sR diodes Vo of Schottky (barrierESR C O ) as freewheeling rectifiers Use 1 PWM = Creduces sdiode(1reverse Q + s 2 / 2input current spikes, = (1 + / ) + s / recovery ) 8 Vc 2 F R p n n P1 7 easing high-side current sensing in the SC4524B. These GPWM R , GCA RS p 1 , RC O Z = 1 11 , R ESRC O
DV = DI ESR +
1

Fig.5
Applications Information (Cont.)
diodes should have an average forward current rating at least 2A and a reverse blocking voltage of at least a few volts higher than the input voltage. For switching regulators operating at low duty cycles (i.e. low output voltage to input voltage conversion ratios), it is beneficial to use freewheeling diodes with somewhat higher average current ratings (thus lower forward voltages). This is because the diode conduction interval is much longer than that of the transistor. Converter efficiency will be improved if the voltage drop across the diode is lower. The freewheeling diode should be placed close to the SW pin of the SC4524B to minimize ringing due to trace inductance. 10BQ015, 20BQ030 (International Rectifier), B220A (Diodes Inc.), SS13, SS22 (Vishay), CMSH1-20M, CMSH1-20ML and CMSH2-20M (Central-Semi.) are all suitable. The freewheeling diode should be placed close to the SW pin of the SC4524B on the PCB to minimize ringing due to trace inductance. Bootstrapping the Power Transistor The minimum BST-SW voltage required to fully saturate the power transistor is shown in Figure 5, which is about 1.96V at room temperature. The BST-SW voltage is supplied by a bootstrap circuit powered from either the input or the output of the BST C1 converter (Figure 6). To maximize efficiency, tie the bootstrap diode toVIN the converter output if VO>2.5V. VOUT SW IN Since the bootstrap supply current is proportional to the SC4524B D converter load current, using a lower voltage2 to power GND the bootstrap circuit reduces driving loss and improves efficiency.
(a) D1 D1 VIN IN
SC4524B
SS270 REV 6-7
2.2 2.1 2.0 1.9 1.8 1.7 1.6 -50
Minimum Bootstrap Voltage vs Temperature
Voltage (V)
ISW = -2.6A
-25
0
25
50
o
75
100 125
Temperature ( C)
Figure 5. Typical Minimum Bootstrap Voltage required to Saturate Transistor (ISW= -2.6A).
D1
BST
C1 SW VOUT VIN
SC4524B
GND
D 2
(a)
BST VIN
C1 SW VOUT
IN
SC4524B
GND
D 2
(b)
For the bootstrap circuit, a fast switching PN diode (such as 1N4148 or 1N914) and a small (0.1F - 0.47F) ceramic capacitor is sufficient for most applications. When bootstrapping from 2.5V to 3.0V output voltages, use a low forward drop Schottky diode (BAT-54 or similar) for D1.
Figure 6. Methods of Bootstrapping the SC4524B
Loop Compensation The goal of compensation is to shape the frequency response of the converter so as to achieve high DC accuracy and fast transient response while maintaining loop stability.
12
SC4524B
Applications Information (Cont.)
CONTROLLER AND SCHOTTKY DIODE Io
CA
Rs
REF
Including the voltage divider (R4 and R6), the control to feedback transfer function is found and plotted in Figure 8 as the converter gain.
SW L1 Vo
+ EA Vc
Vramp
FB
-
PWM MODULATOR
COMP C5 R7 C8 Co
R4
Resr
R6
Figure 7. Block diagram of control loops
1 VFB 1 1 A C diagram in 1 VFB The block= - 20 logFigure 7shows the control loops of a A C = - 20 log G R 2F C V CA S CO O G CA R S 2FC C O innerloop (current VO buck converter with the SC4524B. The loop) consists of a current sensing resistor (Rs=6.1mW) 1 1 1 1 A C = - amplifier VFB and a current 20 log (CA) with gain (GCA=28). The outer -6 A C = - 20 log -3 3 2 80 10 3 22 10 28 6. of an 2 80 10 3 22 a -6 loop (voltage loop) consists1 10 - error amplifier (EA), 10 28 6.1 10 VO PWM modulator, and a LC filter. 1 Since R71= 10 1.0 = 15.9dB current 1 3 the = 10 loopis internally closed, the remaining 22 3k V -6 2 10 R7the0loop 10 - 3 = 22.3k is to design the voltage 0.28 compensation .28 3. - log 80task for22 10 FB1033 G CA R S 2FC C O (CV,OR, and C ). 1 compensator 5 7 1 8 C5 = = 0.45nF C5 = = 0.45nF 3 2 16 10 22.1 10 3 2 16 10 3 122.1 10 3 1.0 1 = F output log For a converter with switching frequency 15.,9dB SW 3 1 6.1 10 -= , output capacitance C 3 12pF 1 28inductance L 2 80 10 3 22 10 -6 = .3 loading R, the and pF C8 = 1 C8 3 3O = 12 3 2 600 10 3 transfer function in Figure 7 is control (VC) 2 output (VO) 22.1 10 to 600 10 22.1 10 0.45nF given by: 15.9 = 22.3kV G (1 + sRESR C O ) GPWM (1 + sRESR C O ) o Vo = 10 -3pF = 12 = (1 + s / PWM + s / Q + s2 / 2 ) 2 2 Vc p )(1 n n Vc (1 + s / p )(1 + s / n Q + s / n ) 1 = 0.45nF 6 10 3 This1 10 3 function has a finite DC gain 22. transfer R 1 1 GPWM R , p 1 , GPWM 3= 12pF , p RC , 2 G R RC O / 2) 0010 3 22.1 10 GCA R S
n CA S O
20 15.9 15.9 20 20
Since the converter gain has only one dominant pole at low frequency, a simple Type-2 compensation network is sufficient for voltage loop compensation. As shown in Figure 8, the voltage compensator has a low frequency integrator pole, a zero at FZ1, and a high frequency pole at FP1. The integrator is used to boost the gain at low frequency. The zero is introduced to compensate the excessive phase lag at the loop gain crossover due to the integrator pole (-90deg) and the dominant pole (-90deg). The high frequency pole nulls the ESR zero and attenuates high frequency noise.
60 1.0 1.0 = 15.9dB 3.3 = 15.9dB 3 .3 30 GAIN (dB) Fz1 Fp1
CO MP EN SA TO RG AIN
0
Fp
Fc CO NV ER TER GA IN
LO OP GA IN
-30 Fz -60 1K Fsw/2
10K
100K FREQUENCY (Hz)
1M
10M
(8)
Figure 8. Bode plots for voltage loop design Therefore, the procedure of the voltage loop design for 1 the SC4524B can be summarized as: Z = 1 , Z = R C , ESR O R ESRC O (1) Plot the converter gain, i.e. control to feedback transfer function. (2) Select the open loop crossover frequency, FC, between 10% and 20% of the switching frequency. At FC, find the required compensator gain, AC. In typical applications with ceramic output capacitors, the ESR zero is neglected and the required compensator gain at FC can be estimated by
AC an ESR zero FZ at AC 20 10 20 101 R= 1 PWM (1 + sRESR7C= ) R7 = g G, Z O g m , m CO R / 2 / p )(1 + s / n Q + s 2ESRCnO) 1 1 C5 = C 5 = low-frequency pole FP at a dominant 2 F R Z1 7 2 FZ1 R7 1 1 , p , Z = , 1 C 8 = RC O1 = RS R ESRC O C 8 2 F R P1 7 2 FP1 R7 and double poles at half the switching frequency. V R4 = R6 O - 1 1.0 V
1 VFB 1 A C = - 20 log G R 2F C V CO O CA S
(9)
1 1 1.0 13 A C = - 20 log -3 3 -6 3. 2 80 10 22 10 28 6.1 10
C5 = C8 =
1 2 16 10 22.1 10 3
3
= 0.45nF
SC4524B
1 = 12pF 2 600 10 3 22.1 10 3
Applications Information (Cont.)
GPWM (1 + sRESR C O ) Vo capacitor, the main power switch and the freewheeling (3) Place = compensator zero, FZ1,2between 10% and the 2 Vc (1 + s / p )(1 + s / n Q + s / n ) diode carry pulse current (Figure 9). For jitter-free 20% of the crossover frequency, FC. operation, the size of the loop formed by these components (4) Use the compensator pole, FP1, to cancel the ESR zero, FZ. R 1 1 should be minimized. Since the power switch is already GPWM , p , Z = , (5) Then, the parameters of the compensation network C GCA RS RC O R ESR integrated within the SC4524B, connecting the anode of O the freewheeling diode close to the negative terminal of can be calculated by AC the input bypass capacitor minimizes size of the switched 10 20 current loop. The input bypass capacitor should be placed R7 = gm close to the IN pin. Shortening the traces of the SW and BST nodes reduces the parasitic trace inductance at these 1 C5 = nodes. This not only reduces EMI but also decreases 2 FZ1 R7 switching voltage spikes at these nodes. 1 C8 = 2 FP1 R7 The exposed pad should be soldered to a large ground plane as the ground copper acts as a heat sink for the where gm=0.28mA/V is the EA gain of the SC4524B. device. To ensure proper adhesion to the ground plane, avoid using vias directly under the device. Example: Determine the voltage compensator for an 800kHz, 12V to 3.3V/2A converter with 22uF ceramic output capacitor.
V IN
Choose a loop gain crossover frequency of 80kHz, and place voltage compensator zero 1 and pole FZ1=16kHz 1 VFB at A C F ), 20 log1 = - and F =600kHz. From Equation (9), the (20% of C 1 V A C = - 20 log P1 G CA R S 2FC C O VO F C VFB G R 2 required compensatorCgain atFC is O O CA S
VOUT
11 1 1.0 1 A C = -A C = - 20 log - 3 20 log = 15.9dB 80 10 3 10 3 3 28 6.1 1028 2 .1 103- 22 2 -680 . 10 3 22 10 - 6 6
Then the compensator parameters are
R7 = 10 20 15.9 = 22.3k 0.28 10 -3 10 20 R7 = 1 = 22.3k 0.28 10 -3 3 = 0.45nF C5 = 3 2 16 10 22.1 10 C8 =
15.9
1.0 = 15.9dB 3 .3
ZL
1 1 = 0.45nF 2 3 16 .1 10 3 = 12.pF 10 3 22 10 3 22 1 2 600 10 C5 = 1 = 12pF
Figure 9. Heavy lines indicate the critical pulse current loop. The inductance of this loop should be minimized.
Select R7=22.1k, C5=0.47nF, and C8=10pF for the design.
1 GPWM (1 + ,sRESR C O )Z = 1 , p RCfor various 2 R ESRC O applications parameters typical 2 Vc (1 + s / p )(1 + O / n Q + s / n ) s are listed in Table 4. A MathCAD program is also available upon10 R7 = request for detailed calculation of the compensator gm parameters. R 1 1 GPWM , p , Z = , 1 C5 = RC O R ESRC O 2 FZ1 R7 GCA RS PCB Layout Considerations R GPWM Vo , GCA=RS Compensator
AC 20
C 8 = (1 + sR C )3 GPWM Vo ESR O 2 600 10 22.1 10 3 = 2 Vc (1 + s / p )(1 + s / n Q + s 2 / n )
Vin
Cu
C8 =
In a step-down switching regulator, the input bypass R7 = gm
1 AC 2 FP1 R7 20 10
1 C5 = 2 F R
+
14
SC4524B
Recommended Component Parameters in Typical Applications
Table 4 lists the recommended inductance (L1) and compensation network (R7, C5, C8) for common input and output voltages. The inductance is determined by assuming that the ripple current is 35% of load current IO. The compensator parameters are calculated by assuming a 22mF low ESR ceramic output capacitor and a loop gain crossover frequency of FSW/10. Table 4. Recommended inductance (L1) and compensator (R7, C5, C8)
Vin(V) Typical Applications Vo(V) Io(A) Fsw(kHz) 500 1 1000 1.5 500 2 1000 500 1 1000 2.5 500 2 1000 500 1 1000 1.5 500 2 1000 500 1 1000 2.5 500 2 1000 500 1 1000 3.3 500 2 1000 1 500 1.5 2 500 500 1 1000 2.5 500 2 1000 500 1 1000 3.3 500 2 1000 500 1 1000 5 500 2 1000 500 1 1000 7.5 500 2 1000 500 1 1000 10 500 2 1000 C2(uF) Recommended Parameters L1(uH) R7(k) C5(nF) C8(pF) 6.8 6.65 2.2 3.3 12.4 0.68 3.3 6.65 2.2 1.5 12.4 0.68 4.7 22.1 0.68 2.2 35.7 0.47 2.2 18.2 0.68 1.5 35.7 0.47 6.8 6.65 2.2 3.3 12.7 0.68 3.3 6.65 2.2 2.2 12.7 0.68 8.2 11.3 1.5 4.7 23.7 0.47 4.7 11.3 1 2.2 20 0.47 6.8 15 0.82 3.3 26.7 0.47 3.3 15 0.82 2.2 29.4 0.47 8.2 7.15 2.2 10 4.7 7.15 2.2 15 11.3 1 6.8 20 0.68 6.8 11.3 1 3.3 20 0.47 15 15 0.82 8.2 30.9 0.47 8.2 15 0.82 4.7 30.9 0.47 15 23.7 0.68 10 41.2 0.47 8.2 23.7 0.68 4.7 45.3 0.47 15 35.7 0.68 8.2 63.4 0.47 8.2 35.7 0.68 4.7 63.4 0.47 10 42.2 0.68 4.7 84.5 0.47 4.7 42.2 0.68 2.2 84.5 0.47
3.3
5
22
12
15
SC4524B
Typical Application Schematics
V IN 5V C4 4.7mF D1 1N4148 C1 0.1mF L1 2.2mH R4 33.2k
IN
BST SW
OUT 3.3V/2A
SS/EN
SC4524B
FB
COMP C7 10nF R7 29.4k
ROSC
GND D2 20BQ030 R6 14.3k C2 22mF
C8 10pF
R5 18.2k
C5 0.47nF
L1: Coiltronics LD1-2R2
C2: Murata GRM31CR60J226K C4: Murata GRM31CR60J475K
Figure 10. 1MHz 5V to 3.3V/2A Step-down Converter
V
IN
10V - 16V C4 4.7mF
D1 1N4148 IN BST SW
C1 0.33mF L1 4.7mH R4 33.2k
OUT
SS/EN
SC4524B
FB
EVB #a
1.5V/2A C2 22mF
COMP C7 10nF R7 7.15k
ROSC
GND D2 20BQ030 R6 66.5k
C8 10pF
R5 47.3k
C5 2.2nF
L1: Coiltronics DR73-4R7
C2: Murata GRM31CR60J226K C4: Murata GRM31CR60J475K
Figure 11. 500kHz 10V-16V to 1.5V/2A Step-down Converter
EVB #b
16
SC4524B
SS
Typical Performance Characteristics
SS270 REV 6-7
(For A 12V to 5V/2A Step-down Converter with 1MHz Switching Frequency)
Load Characteristic
6 5
Output Voltage (V)
4 3 2 1 0 0 0.5 1 1.5 2 2.5 3
12V Input (5V/DIV)
5V Output (2V/DIV)
SS Voltage (1V/DIV)
Load Current (A)
10ms/DIV
Figure 12(a). Load Characteristic
OCP
Figure 12(b). VIN Start up Transient (IO=2A)
5V Output Short (5V/DIV)
5V Output Response (500mV/DIV, AC Coupling)
Inductor Current (1A/DIV)
Retry Inductor Current (2A/DIV)
SS Voltage (2V/DIV)
40us/DIV
20ms/DIV
Figure 12(c). Load Transient Response (IO= 0.3A to 2A)
Figure 12(d). Output Short Circuit (Hiccup)
17
SC4524B
SO-8 EDP2 Outline
Outline Drawing - SOIC-8 EDP
A N 2X E/2 E1 E 1 ccc C 2X N/2 TIPS 2 e/2 B D aaa C SEATING PLANE A2 A A1 C A-B D e
D
DIM
A A1 A2 b c D E1 E e F H h L L1 N 01 aaa bbb ccc
DIMENSIONS INCHES MILLIMETERS MIN NOM MAX MIN NOM MAX
.053 .069 .000 .005 .049 .065 .012 .020 .007 .010 .189 .193 .197 .150 .154 .157 .236 BSC .050 BSC .116 .120 .130 .085 .095 .099 .010 .020 .016 .028 .041 (.041) 8 0 8 .004 .010 .008 1.35 1.75 0.00 0.13 1.25 1.65 0.31 0.51 0.17 0.25 4.80 4.90 5.00 3.80 3.90 4.00 6.00 BSC 1.27 BSC 2.95 3.05 3.30 2.15 2.41 2.51 0.25 0.50 0.40 0.72 1.04 (1.05) 8 0 8 0.10 0.25 0.20
C
bxN bbb F
h
EXPOSED PAD H
H GAGE PLANE 0.25
h
c
L (L1)
01
SEE DETAIL SIDE VIEW
NOTES: 1.
A
DETAIL
A
CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. DATUMS -A- AND -B- TO BE DETERMINED AT DATUM PLANE -H3. DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. 4. REFERENCE JEDEC STD MS-012, VARIATION BA.
Land Pattern - SOIC-8 EDP
SO-8 EDP2 Landing Pattern
E D
SOLDER MASK
DIM
(C) F G Z C D E F G P X Y Z
DIMENSIONS INCHES MILLIMETERS
(.205) .134 .201 .101 .118 .050 .024 .087 .291 (5.20) 3.40 5.10 2.56 3.00 1.27 0.60 2.20 7.40
Y THERMAL VIA O 0.36mm
NOTES:
P X
1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR COMPANY'S MANUFACTURING GUIDELINES ARE MET. 2. REFERENCE IPC-SM-782A, RLP NO. 300A. 3. THERMAL VIAS IN THE LAND PATTERN OF THE EXPOSED PAD SHALL BE CONNECTED TO A SYSTEM GROUND PLANE. FAILURE TO DO SO MAY COMPROMISE THE THERMAL AND/OR FUNCTIONAL PERFORMANCE OF THE DEVICE.
Contact Information
Semtech Corporation Power Mangement Products Division 200 Flynn Road, Camarillo, CA 93012 Phone: (805) 498-2111 Fax: (805) 498-3804
18

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